Certain types of voice/data receivers are connected to transmitters through transmission lines which may be the lines of the commercial telephone network. One of the parameters of a transmission line of this type is its characteristic impedance, which is the impedance that would be measured at the end of such a line if it were infinitely long. The importance of this characteristic impedance lies in the fact that if any length of line is terminated in an impedance of this value, all the energy flowing along the line is absorbed at the termination and none is reflected back along the line to interfere with the transmitted information. A result of this is that the input impedance of any length of transmission line terminated in its characteristic impedance is equal to its characteristic impedance.
In prior art receivers of this type, an incoming transmission line is connected to the primary winding of an input transformer which has, across its secondary winding, a load resistor whose value is nearly as high as the characteristic impedance of the line. For example, if the characteristic impedance of the line is 600 ohms, the load resistor connected across the secondary winding of the input transformer is about 600 ohms minus the transformer impedance, such that the combined impedance of the primary and secondary windings of the transformer and of the load resistor is approximately 600 ohms. A typical example of a prior art arrangement of this type is illustrated in FIG. 1 where an incoming transmission line is connected to the primary winding of a transformer T, this primary winding having an impedance made up of resistance TR1 of approximately 30 ohms and reactance L1. The secondary winding is similarly represented as resistive TR2 of about 30 ohms and reactance L2. The secondary winding has a load resistor R3 of about 540 ohms across it. For the relevant frequencies, i.e., in or about the audio range, the impedance in which the transmission line is terminated is approximately the sum of the resistive impedance components of the primary and secondary windings of the transformer T and the resistor R3, i.e., about 600 ohms, which is a proper match for a transmission line having a characteristic impedance of 600 ohms.
However, this proper match becomes a disadvantage when low frequency signals are being sent to transformer T over the transmission line. For example, when the signal is a rectangular positive voltage pulse of some width, there is a rise of the current through and voltage across R3 corresponding to the rise of this pulse, but then that current and voltage start decaying or "sagging" because the impedance of the secondary winding starts becoming very low as compared to the high value of R3 in the prior art example shown in FIG. 1. Stated differently, after the initial rise in the pulse, the secondary winding of T starts shorting out R3.
One measure of the low frequency response of networks of this type is the ratio of the resistance at the secondary winding of the input transformer to the inductance of the transformer. In the case of the FIG. 1 arrangement, this is the ratio of about 570 ohms (TR2 + R3) to the inductance of the input transformer, which may be, for example, about 3 to 4 Henry. This ratio is therefore in the range of about 190 to 140. The lower this ratio is the better the low frequency response of signals transmitted through the input transformer. In qualitative terms, this means that when this ratio is high, the sag discussed above is high and the height of a rectangular positive pulse is not maintained at R3. Conversely, when this ratio is low, the sag is low and the height of such a pulse is better maintained at R3. However, in the prior art this large load resistor R3 (causing a high ratio) was considered a necessity required by the constraint that the characteristic impedance of the line must be matched by the network terminating the line. The conflict had to be resolved in favor of a matched line termination because otherwise a reflected wave would be passed back onto the line and would interfere with the information signal being transmitted to the input transformer and perhaps with other information signals sharing the line. It is not practical to lower the ratio by increasing the inductance of the input transformer because of the unacceptable size and cost penalties that would ensue.
In accordance with the invention, and in contradiction to conventional design rules, an input transformer in an arrangement of this type works into substantially short circuit impedance. Despite this seemingly improper arrangement, the incoming transmission line is terminated into a proper impedance because suitable termination resistors are connected in series with the primary winding of the input transformer. The advantage of this seemingly improper arrangement is that the load resistance into which the input transformer works is very low, permitting the transformer coupling to have considerably improved low frequency response and less sag. In qualitative terms, the signal now faces the combination of the same inherent resistance of the secondary winding of the input transformer (provided the same transformer is used) but a much smaller load resistor. This smaller load resistor draws a greater proportion of the current at the secondary side of T as compared with R3 in FIG. 1, and tends to maintain a current flow and a potential difference with less sag. For example, using the same input transformer as in FIG. 1 but terminating its secondary winding in an actual or virtual resistance of about 10 ohms, as contrasted with 540 ohms in the FIG. 1 arrangement, and assuming that the secondary winding has an inherent resistance of 30 ohms and that the transformer inductance is 3 Henry, the frequency response measure for FIG. 1 would be about 190 while the same measure would be about 13 in an arrangement in accordance with the invention. This is an improvement in low frequency response by a factor of about 14 or 15 when the above-identified measure of low frequency response is used.
Referring to FIG. 1a for a qualitative illustration, the waveform 14' is the beginning of a wide positive pulse, the waveform 16' is the ideal response across R3, the waveform 18 is the response across a small load resistor in accordance with this invention, and the waveform 20 is the response across R3 in the prior art circuit of FIG. 1. In FIG. 1a the axis pointing up is voltage and the axis pointing to the right is time. This considerable improvement in low frequency response is present even if the input transformer is identical to the input transformer used in the corresponding prior art arrangements. Moreover, it is possible to use a substantially smaller, and therefor less costly and less bulky, input transformer and still retain the favorable characteristics resulting from using the invention. For example, if a less costly and less bulky transformer is used, e.g., with an inductance of only 1 Henry, everything else being the same, the low frequency response measure for an arrangement in accordance with the invention would be about 40, and if a still less costly and less bulky transformer is used, with a 0.3 Henry inductance, the low frequency response measure would be about 133, still being better than the 140 to 190 measure of the prior art arrangement of FIG. 1.